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19-0881; Rev 2; 1/10
KIT ATION EVALU BLE AVAILA
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
General Description Features
Up to 30A Output Current True-Differential Remote Output Sensing Average Current-Mode Control 4.75V to 5.5V or 7V to 28V Input-Voltage Range 0.1V/0.03V LED Current-Sense Options Maximize Efficiency (MAX16821B/MAX16821C) Thermal Shutdown Nonlatching Output Overvoltage Protection Low-Side Buck Mode with or without Synchronous Rectification High-Side Buck and Low-Side Boost Mode with or without Synchronous Rectification 125kHz to 1.5MHz Programmable/Synchronizable Switching Frequency Integrated 4A Gate Drivers Clock Output for 180 Out-of-Phase Operation for Second Driver -40C to +125C Operating Temperature Range
MAX16821A/MAX16821B/MAX16821C
The MAX16821A/MAX16821B/MAX16821C pulsewidth-modulation (PWM) LED driver controllers provide high output-current capability in a compact package with a minimum number of external components. The MAX16821A/MAX16821B/MAX16821C are suitable for use in synchronous and nonsynchronous step-down (buck), boost, buck-boost, SEPIC, and Cuk LED drivers. A logic input (MODE) allows the devices to switch between synchronous buck and boost modes of operation. These devices are the first high-power drivers designed specifically to accommodate common-anode HBLEDs. The ICs offer average current-mode control that enable the use of MOSFETs with optimal charge and on-resistance figure of merit, thus minimizing the need for external heatsinking even when delivering up to 30A of LED current. The differential sensing scheme provides accurate control of the LED current. The ICs operate from a 4.75V to 5.5V supply range with the internal regulator disabled (VCC connected to IN). These devices operate from a 7V to 28V input supply voltage with the internal regulator enabled. The MAX16821A/MAX16821B/MAX16821C feature a clock output with 180 phase delay to control a second out-of-phase LED driver to reduce input and output filter capacitor size and to minimize ripple currents. The wide switching frequency range (125kHz to 1.5MHz) allows the use of small inductors and capacitors. Additional features include programmable overvoltage protection and an output enable function.
Ordering Information
PART MAX16821AATI+ MAX16821BATI+ TEMP RANGE -40C to +125C -40C to +125C PINPACKAGE 28 TQFN-EP* 28 TQFN-EP*
MAX16821CATI+ -40C to +125C 28 TQFN-EP* +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad.
Simplified Diagram
7V TO 28V
Applications
Front Projectors/Rear Projection TVs Portable and Pocket Projectors Automotive Exterior Lighting LCD TVs and Display Backlight Automotive Emergency Lighting and Signage
I.C. IN EN DH Q1
C1
L1
VLED
MAX16821
DL Q2 C2 Q3 OVI CLP CSP R1 PGND
Typical Operating Circuit and Selector Guide appear at end of data sheet.
. HIGH-FREQUENCY PULSE TRAIN
________________________________________________________________ Maxim Integrated Products
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
ABSOLUTE MAXIMUM RATINGS
IN to SGND.............................................................-0.3V to +30V BST to SGND..........................................................-0.3V to +35V BST to LX..................................................................-0.3V to +6V DH to LX ...........................................-0.3V to (VBST - VLX) + 0.3V DL to PGND................................................-0.3V to (VDD + 0.3V) VCC to SGND............................................................-0.3V to +6V VCC, VDD to PGND ...................................................-0.3V to +6V SGND to PGND .....................................................-0.3V to +0.3V VCC Current ......................................................................300mA All Other Pins to SGND...............................-0.3V to (VCC + 0.3V) Continuous Power Dissipation (TA = +70C) 28-Pin TQFN 5mm x 5mm (derate 34.5mW/C above +70C) ............................................................2758mW Operating Temperature Range .........................-40C to +125C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VCC = 5V, VDD = VCC, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER Input-Voltage Range Quiescent Supply Current LED CURRENT REGULATOR VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821A) VIN = 7V to 28V, fSW = 500kHz (MAX16821A) VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821B) VIN = 7V to 28V, fSW = 500kHz (MAX16821B) VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821C) VIN = 7V to 28V, fSW = 500kHz (MAX16821C) Soft-Start Time STARTUP/INTERNAL REGULATOR VCC Undervoltage Lockout (UVLO) UVLO Hysteresis VCC Output Voltage MOSFET DRIVER Output Driver Impedance Output Driver Source/Sink Current Nonoverlap Time IDH, IDL tNO CDH/DL = 5nF Low or high output, ISOURCE/SINK = 20mA 1.1 4 35 3 A ns UVLO VCC rising VCC falling VIN = 7V to 28V, ISOURCE = 0 to 60mA 4.85 4.1 4.3 200 5.10 5.30 4.5 V mV V tSS 0.594 0.594 0.098 0.098 0.028 0.028 0.600 0.600 0.100 0.100 0.030 0.030 1024 0.606 0.606 0.102 V 0.102 0.032 0.032 Clock Cycles SYMBOL VIN IQ CONDITIONS Internal LDO on Internal LDO off (VCC connected to VIN) VEN = VCC or SGND, no switching MIN 7 4.75 2.7 TYP MAX 28 5.50 5.5 UNITS V mA
Differential Set Value (VSENSE+ to VSENSE-) (Note 2)
2
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER OSCILLATOR Switching Frequency Range RT = 500k Switching Frequency fSW RT = 120k RT = 39.9k Switching Frequency Accuracy CLKOUT Phase Shift with Respect to DH (Rising Edges) CLKOUT Phase Shift with Respect to DL (Rising Edges) CLKOUT Output-Voltage Low CLKOUT Output-Voltage High SYNC Input High Pulse Width SYNC Input Clock High Threshold SYNC Input Clock Low Threshold SYNC Pullup Current SYNC Power-Off Level INDUCTOR CURRENT LIMIT Average Current-Limit Threshold Reverse Current-Limit Threshold Cycle-by-Cycle Current Limit Cycle-by-Cycle Overload CURRENT-SENSE AMPLIFIER CSP to CSN Input Resistance Common-Mode Range Input Offset Voltage Amplifier Voltage Gain 3dB Bandwidth Transconductance Open-Loop Gain RCS VCMR(CS) VOS(CS) AV(CS) f3dB gm AVL(CE) VIN = 7V to 28V 0 0.1 34.5 4 550 50 4 5.5 k V mV V/V MHz S dB VCL VCLR CSP to CSN CSP to CSN CSP to CSN VCSP to VCSN = 75mV 26.4 27.5 -2.0 60 260 33.0 mV mV mV ns VOL VOH tSYNC VSYNCH VSYNCL ISYNC_OUT VSYNC_OFF VRT/SYNC = 0V 250 120k < RT 500k 40k RT 120k fSW = 125kHz, MODE connected to SGND fSW = 125kHz, MODE connected to VCC ISINK = 2mA ISOURCE = 2mA 4.5 200 2 0.4 500 0.4 125 120 495 1515 -5 -8 180 Degrees 180 0.4 V V ns V V A V 125 521 1620 1500 130 547 1725 +5 +8 % kHz kHz SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX16821A/MAX16821B/MAX16821C
CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER Common-Mode Voltage Range DIFF Output Voltage Input Offset Voltage SYMBOL VCMR(DIFF) VCM VOS(DIFF) VSENSE+ = VSENSE- = 0V MAX16821A MAX16821B/MAX16821C MAX16821A Amplifier Voltage Gain AV(DIFF) MAX16821B MAX16821C MAX16821A, CDIFF = 20pF 3dB Bandwidth f3dB MAX16821B, CDIFF = 20pF MAX16821C, CDIFF = 20pF MAX16821A SENSE+ to SENSE- Input Resistance OUTV AMPLIFIER Gain-Bandwidth Product 3dB Bandwidth Output Sink Current Output Source Current Maximum Load Capacitance OUTV to (CSP - CSN) Transfer Function Input Offset Voltage VOLTAGE-ERROR AMPLIFIER (EAOUT) Open-Loop Gain Unity-Gain Bandwidth EAN Input Bias Current Error Amplifier Output Clamping Voltage INPUTS (MODE AND OVI) MODE Input-Voltage High MODE Input-Voltage Low MODE Pulldown Current OVI Trip Threshold OVI Hysteresis OVI Input Bias Current OVPTH OVIHYS IOVI VOVI = 1V 4 1.244 5 1.276 200 0.2 2 0.8 6 1.308 V V A V mV A AVOLEA fGBW IB(EA) VCLAMP(EA) VEAN = 2V With respect to VCM -0.2 905 70 3 +0.03 930 +0.2 940 dB MHz A mV 4mV CSP - CSN 32mV 132.5 VOUTV = 2V VOUTV = 2V 30 80 50 135 1 137.7 4 1 MHz MHz A A pF V/V mV RVS MAX16821B MAX16821C 50 30 10 -3.7 -1.5 0.992 5.85 18.5 1 6 20 1.7 1600 550 100 60 20 k CONDITIONS MIN 0 0.6 +3.7 +1.5 1.008 6.1 21.5 MHz kHz V/V TYP MAX 1.0 UNITS V V mV
LED CURRENT SIGNAL DIFFERENTIAL VOLTAGE AMPLIFIER (DIFF)
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C.) (Note 1)
PARAMETER ENABLE INPUT (EN) EN Input-Voltage High EN Input Hysteresis EN Pullup Current THERMAL SHUTDOWN Thermal Shutdown Thermal-Shutdown Hysteresis 165 20 C C IEN 13.5 EN rising 2.437 2.5 0.28 15 16.5 2.562 V V A SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX16821A/MAX16821B/MAX16821C
Note 1: All devices are 100% production tested at +25C. Limits over temperature are guaranteed by design. Note 2: Does not include an error due to finite error amplifier gain. See the Voltage-Error Amplifier section.
Typical Operating Characteristics
(VIN = 12V, VDD = VCC = 5V, TA = +25C, unless otherwise noted.)
SUPPLY CURRENT (IQ) vs. FREQUENCY
MAX16821A toc01
SUPPLY CURRENT vs. TEMPERATURE
MAX16821A toc02
VCC LOAD REGULATION vs. VIN
5.4 5.3 5.2 VCC (V) 5.1 5.0 4.9 VIN = 24V VIN = 12V
MAX16821A toc03
10 9 8 SUPPLY CURRENT (mA) 7 6 5 4 3 2 1 0 100 300 500 700 VIN = 5V VIN = 12V VIN = 24V EXTERNAL CLOCK NO DRIVER LOAD
70 65 SUPPLY CURRENT (mA) 60 55 50 45 40
5.5
4.8 4.7 VIN = 12V CDH/DL = 22nF -40 -15 10 35 60 85 4.6 4.5 0
VIN = 7V
900 1100 1300 1500
15 30 45 60 75 90 105 120 135 150 VCC LOAD CURRENT (mA)
FREQUENCY (kHz)
TEMPERATURE (C)
DRIVER RISE TIME vs. DRIVER LOAD CAPACITANCE
MAX16821A toc04
DRIVER FALL TIME vs. DRIVER LOAD CAPACITANCE
MAX16821A toc05
HIGH-SIDE DRIVER (DH) SINK AND SOURCE CURRENT
MAX16821A toc06
200 180 160 140 tR (ns)
100
CLOAD = 22nF VIN = 12V
80
100 80 60 40 20 0 0 5 10 15 20 25 LOAD CAPACITANCE (nF) DL DH
fF (ns)
120
60 2A/div 40 DL DH 20
0 0 5 10 15 20 25 100ns/div LOAD CAPACITANCE (nF)
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Typical Operating Characteristics (continued)
(VIN = 12V, VDD = VCC = 5V, TA = +25C, unless otherwise noted.)
LOW-SIDE DRIVER (DL) SINK AND SOURCE CURRENT
MAX16821A toc07
HIGH-SIDE DRIVER (DH) RISE TIME
MAX16821A toc08
HIGH-SIDE DRIVER (DH) FALL TIME
MAX16821A toc09
CLOAD = 22nF VIN = 12V
VIN = 12V DH RISING
CLOAD = 22nF VIN = 12V
3A/div
2V/div
2V/div
100ns/div
40ns/div
40ns/div
LOW-SIDE DRIVER (DL) RISE TIME
MAX16821A toc10
LOW-SIDE DRIVER (DL) FALL TIME
MAX16821A toc11
FREQUENCY vs. RT
VIN = 12V
MAX16821A toc12
10,000
CLOAD = 22nF VIN = 12V
CLOAD = 22nF VIN = 12V
2V/div
2V/div
fSW (kHz)
1000
100 40ns/div 40ns/div 30 70 110 150 190 230 270 310 350 390 430 470 510 550 RT (k)
FREQUENCY vs. TEMPERATURE
258 256 254 fSW (kHz) 252 250 248 246 244 242 240 0 5 10 15 20 25 30 35 VIN = 12V
MAX16821A toc13
SYNC, CLKOUT, AND DH WAVEFORMS
MAX16821A toc14
SYNC, CLKOUT, AND DL WAVEFORMS
MAX16821A toc15
260
RT/SYNC 5V/div 0V MODE = SGND CLKOUT 5V/div 0V DH 5V/div 0V 1s/div 1s/div MODE = VCC
RT/SYNC 5V/div 0V
CLKOUT 5V/div 0V DL 5V/div 0V
TEMPERATURE (C)
6
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
Pin Description
PIN 1 2, 7 3 4 5 6 8, 22, 25 9 10 11 12 13 14 15 16 17 18 19 20 21 23 24 26 27 28 -- NAME PGND N.C. DL BST LX DH SGND CLKOUT MODE EN RT/SYNC OUTV I.C. OVI CLP EAOUT EAN DIFF CSN CSP SENSESENSE+ IN VCC VDD EP Power-Supply Ground No Connection. Not internally connected. Low-Side Gate-Driver Output Boost-Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver supply. Connect a ceramic capacitor between BST and LX. High-Side MOSFET Source Connection High-Side Gate-Driver Output Signal Ground. SGND is the ground connection for the internal control circuitry. Connect SGND and PGND together at one point near the IC. Oscillator Output. If MODE is low, the rising edge of CLKOUT phase shifts from the rising edge of DH by 180. If MODE is high, the rising edge of CLKOUT phase shifts from the rising edge of DL by 180. Buck/Boost Mode Selection Input. Drive MODE low for low-side buck mode operation. Drive MODE high for boost or high-side buck mode operation. MODE has an internal 5A pulldown current to ground. Output Enable. Drives EN high or leave unconnected for normal operation. Drive EN low to shut down the power drivers. EN has an internal 15A pullup current. Switching Frequency Programming. Connect a resistor from RT/SYNC to SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency with an external clock. Inductor Current-Sense Output. OUTV is an amplifier output voltage proportional to the inductor current. The voltage at OUTV = 135 x (VCSP - VCSN). Internally Connected. Connect to SGND for proper operation. Overvoltage Protection. When OVI exceeds the programmed output voltage by 12.7%, the low-side and the high-side drivers are turned off. When OVI falls 20% below the programmed output voltage, the drivers are turned on after power-on reset and soft-start cycles are completed. Current-Error-Amplifier Output. Compensate the current loop by connecting an RC network to ground. Voltage-Error-Amplifier Output. Connect EAOUT to the external gain-setting network. Voltage-Error-Amplifier Inverting Input Differential Remote-Sense Amplifier Output. DIFF is the output of a precision amplifier with SENSE+ and SENSE- as inputs. Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is amplified internally by the current-sense amplifier (Gain = 34.5) to measure the inductor current. Current-Sense Differential Amplifier Positive Input. The differential voltage between CSP and CSN is amplified internally by the current-sense amplifier (Gain = 34.5) to measure the inductor current. Differential LED Current-Sensing Negative Input. Connect SENSE- to the negative side of the LED currentsense resistor or to the negative feedback point. Differential LED Current-Sensing Positive Input. Connect SENSE+ to the positive side of the LED currentsense resistor, or to the positive feedback point. Supply Voltage Input. Connect IN to VCC, for a 4.75V to 5.5V input supply range. Internal +5V Regulator Output. VCC is derived from VIN. Bypass VCC to SGND with 4.7F and 0.1F ceramic capacitors. Low-Side Driver Supply Voltage Exposed Pad. EP is internally connected to SGND. Connect EP to a large-area ground plane for effective power dissipation. Connect EP to SGND. Do not use as a ground connection. FUNCTION
MAX16821A/MAX16821B/MAX16821C
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Detailed Description
The MAX16821A/MAX16821B/MAX16821C are high-performance average current-mode PWM controllers for high-power and high-brightness LEDs (HBLEDs). The average current-mode control technique offers inherently stable operation, reduces component derating and size by accurately controlling the inductor current. The devices achieve high efficiency at high currents (up to 30A) with a minimum number of external components. A logic input (MODE) allows the LED driver to switch between buck and boost modes of operation. The MAX16821A/MAX16821B/MAX16821C feature a CLKOUT output 180 out-of-phase with respect to either the high-side or low-side driver, depending on MODE's logic level. CLKOUT provides the drive for a second out-of-phase LED driver for applications requiring reduced input capacitor ripple current while operating another LED driver. The MAX16821A/MAX16821B/MAX16821C consist of an inner average current regulation loop controlled by an outer loop. The combined action of the inner current loop and outer voltage loop corrects the LED current errors by adjusting the inductor current resulting in a tightly regulated LED current. The differential amplifier (SENSE+ and SENSE- inputs) senses the LED current using a resistor in series with the LEDs and produces an amplified version of the sense voltage at DIFF. The resulting amplified sensed voltage is compared against an internal 0.6V reference at the error amplifier input.
Undervoltage Lockout (UVLO)
The MAX16821A/MAX16821B/MAX16821C include UVLO and a 2048 clock-cycle power-on-reset circuit. The UVLO rising threshold is set to 4.3V with 200mV hysteresis. Hysteresis at UVLO eliminates chattering during startup. Most of the internal circuitry, including the oscillator, turns on when the input voltage reaches 4V. The MAX16821A/MAX16821B/MAX16821C draw up to 3.5mA of quiescent current before the input voltage reaches the UVLO threshold.
Soft-Start
The MAX16821A/MAX16821B/MAX16821C include an internal soft-start for a glitch-free rise of the output voltage. After 2048 power-on-reset clock cycles, a 0.6V reference voltage connected to the positive input of the internal error amplifier ramps up to its final value after 1024 clock cycles. Soft-start reduces inrush current and stress on system components. During soft-start, the LED current will ramp monotonically towards its final value.
Internal Oscillator
The internal oscillator generates a clock with the frequency inversely proportional to the value of RT (see the Typical Operating Circuit). The oscillator frequency is adjustable from 125kHz to 1.5MHz range using a single resistor connected from RT/SYNC to SGND. The frequency accuracy avoids the overdesign, size, and cost of passive filter components like inductors and capacitors. Use the following equation to calculate the oscillator frequency: For 120k RT 500k: fSW = 6.25 x 1010 (Hz) RT
Input Voltage
The MAX16821A/MAX16821B/MAX16821C operate with a 4.75V to 5.5V input supply range when the internal LDO is disabled (VCC connected to IN) or a 7V to 28V input supply range when the internal LDO is enabled. For a 7V to 28V input voltage range, the internal LDO provides a regulated 5V output with 60mA of sourcing capability. Bypass VCC to SGND with 4.7F and 0.1F low-ESR ceramic capacitors. The MAX16821A/MAX16821B/MAX16821C's VDD input provides supply voltage for the low-side and the highside MOSFET drivers. Connect VDD to VCC using an R-C filter to isolate the analog circuits from the MOSFET drivers. The internal LDO powers up the MAX16821A/ MAX16821B/MAX16821C. For applications utilizing a 5V input voltage, disable the internal LDO by connecting IN and VCC together. The 5V power source must be in the 4.75V to 5.5V range of for proper operation of the MAX16821A/MAX16821B/MAX16821C.
For 40k RT 120k: fSW = 6.40 x 1010 (Hz) RT
The oscillator also generates a 2VP-P ramp signal for the PWM comparator and a 180 out-of-phase clock signal at CLKOUT to drive a second out-of-phase LED current regulator.
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
VCC
EN 0.5 x VCC IN +5V LDO UVLO POR TEMP SEN TO INTERNAL CIRCUIT
VCC I.C. CLP CSP AV = 34.5 CSN VCM
AV = 4 OUTV
VCLAMP LOW
gm BST VCLAMP HIGH PWM COMPARATOR S Q MUX R Q DH LX VDD DL
CLK RT/SYNC OSCILLATOR
CLKOUT DIFF SENSEDIFF AMP SENSE+ EAOUT EAN
RAMP GENERATOR
2 x fS VTH
PGND
VCM
MODE
ERROR AMP
0.12 x VREF
OVP COMPARATOR
VREF = 0.6V
SOFTSTART
ENABLE UVLO
MAX16821A MAX16821B MAX16821C
VCM
OVI
SGND
Figure 1. Internal Block Diagram
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Synchronization
The MAX16821A/MAX16821B/MAX16821C synchronize to an external clock connected to RT/SYNC. The application of an external clock at RT/SYNC disables the internal oscillator. Once the MAX16821A/MAX16821B/ MAX16821C are synchronized to an external clock, the external clock cannot be removed if reliable operation is to be maintained. MAX16821C outer LED-current control loop consists of a differential amplifier (DIFF), a reference voltage, and a voltage-error amplifier (VEA).
Inductor Current-Sense Amplifier
The differential current-sense amplifier (CSA) provides a 34.5V/V DC gain. The typical input offset voltage of the current-sense amplifier is 0.1mV with a 0 to 5.5V commonmode voltage range (VIN = 7V to 28V). The current-sense amplifier senses the voltage across RS. The maximum common-mode voltage is 3.2V when VIN = 5V.
Control Loop
The MAX16821A/MAX16821B/MAX16821C use an average current-mode control scheme to regulate the output current (Figure 2). The main control loop consists of an inner current regulation loop for controlling the inductor current and an outer current regulation loop for regulating the LED current. The inner current regulation loop absorbs the double pole of the inductor and output capacitor combination reducing the order of the outer current regulation loop to that of a single-pole system. The inner current regulation loop consists of a current-sense resistor (RS), a current-sense amplifier (CSA), a current-error amplifier (CEA), an oscillator providing the carrier ramp, and a PWM comparator (CPWM) (Figure 2). The MAX16821A/MAX16821B/
Inductor Peak-Current Comparator
The peak-current comparator provides a path for fast cycle-by-cycle current limit during extreme fault conditions, such as an inductor malfunction (Figure 3). Note the average current-limit threshold of 27.5mV still limits the output current during short-circuit conditions. To prevent inductor saturation, select an inductor with a saturation current specification greater than the average current limit. The 60mV threshold for triggering the peak-current limit is twice the full-scale average current-limit voltage threshold. The peak-current comparator has only a 260ns delay.
CCP
RCF RIN RF CF CCZ
DIFF
EAN
EAOUT
CSN
CSP
CLP
CA
VIN
SENSE+ DIFF SENSEVEA
CEA CPWM DRIVER
L
LED STRING
COUT VREF MODE = SGND RS RLS
Figure 2. MAX16821A/MAX16821B/MAX16821C Control Loop
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
Current-Error Amplifier
The MAX16821A/MAX16821B/MAX16821C include a transconductance current-error amplifier with a typical gm of 550S and 320A output sink and source capability. The current-error amplifier output (CLP) is connected to the inverting input of the PWM comparator. CLP is also externally accessible to provide frequency compensation for the inner current regulation loop (Figure 2). Compensate CEA so the inductor current negative slope, which becomes the positive slope to the inverting input of the PWM comparator, is less than the slope of the internally generated voltage ramp (see the Compensation section). In applications without synchronous rectification, the LED driver can be turned off and on instantaneously by shorting or opening the CLP to ground. 2VP-P ramp signal. At the start of each clock cycle, an R-S flip-flop resets and the high-side driver (DH) turns on if MODE is connected to SGND, and DL turns on if MODE is connected to VCC. The comparator sets the flip-flop as soon as the ramp signal exceeds the CLP voltage, thus terminating the ON cycle. See Figure 3.
MAX16821A/MAX16821B/MAX16821C
Differential Amplifier
The differential amplifier (DIFF) allows LED current sensing (Figure 2). It provides true-differential LED current sensing, and amplifies the sense voltage by a factor of 1 (MAX16821A), 6 (MAX16821B), and 20 (MAX16821C), while rejecting common-mode voltage errors. The VEA provides the difference between the differential amplifier output (DIFF) and the desired LED current-sense voltage. The differential amplifier has a bandwidth of 1.7MHz (MAX16821A), 1.6MHz (MAX16821B), and 550kHz (MAX16821C). The difference between SENSE+ and SENSE- is regulated to +0.6V (MAX16821A), +0.1V (MAX16821B), or +0.03V (MAX16821C).
PWM Comparator and R-S Flip-Flop
An internal PWM comparator sets the duty cycle by comparing the output of the current-error amplifier to a
60mV
PEAK-CURRENT COMPARATOR
CLP CSP AV = 34.5
CSN gm = 550S PWM COMPARATOR S RAMP CLK R Q MODE = GND BST Q DH LX VDD DL PGND SHDN
IN
Figure 3. MAX16821A/MAX16821B/MAX16821C Phase Circuit
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Voltage-Error Amplifier (VEA)
The VEA sets the gain of the voltage control loop, and determines the error between the differential amplifier output and the internal reference voltage. The VEA output clamps to 0.93V relative to the internal commonmode voltage, VCM (+0.6V), limiting the average maximum current. The maximum average current-limit threshold is equal to the maximum clamp voltage of the VEA divided by the gain (34.5) of the current-sense amplifier. This results in accurate settings for the average maximum current.
Current Limit
The error amplifier (VEA) output is clamped between -0.050V and +0.93V with respect to common-mode voltage (VCM). Average current-mode control limits the average current sourced by the converter during a fault condition. When a fault condition occurs, the VEA output clamps to +0.93V with respect to the commonmode voltage (0.6V) to limit the maximum current sourced by the converter to ILIMIT = 0.0275 / RS.
Overvoltage Protection
The OVP comparator compares the OVI input to the overvoltage threshold. The overvoltage threshold is typically 1.127 times the internal 0.6V reference voltage plus VCM (0.6V). A detected overvoltage event trips the comparator output turning off both high-side and lowside MOSFETs. Add an RC delay to reduce the sensitivity of the overvoltage circuit and avoid unnecessary tripping of the converter (Figure 4). After the OVI voltage falls below 1.076V (typ.), high-side and low-side drivers turn on only after a 2048 clock-cycle POR and a 1024 clock-cycle soft-start have elapsed. Disable the overvoltage function by connecting OVI to SGND.
MOSFET Gate Drivers
The high-side (DH) and low-side (DL) drivers drive the gates of external n-channel MOSFETs. The drivers' 4A peak sink- and source-current capability provides ample drive for the fast rise and fall times of the switching MOSFETs. Faster rise and fall times result in reduced cross-conduction losses. Size the high-side and low-side MOSFETs to handle the peak and RMS currents during overload conditions. The driver block also includes a logic circuit that provides an adaptive nonoverlap time to prevent shoot-through currents during transition. The typical nonoverlap time is 35ns between the high-side and low-side MOSFETs.
BST
The MAX16821A/MAX16821B/MAX16821C provide power to the low-side and high-side MOSFET drivers through VDD. A bootstrap capacitor from BST to LX provides the additional boost voltage necessary for the high-side driver. VDD supplies power internally to the low-side driver. Connect a 0.47F low-ESR ceramic capacitor between BST and LX and a Schottky diode from BST to VDD.
COVI
RA OVI RB VOUT
MAX16821A MAX16821B MAX16821C
DIFF RIN EAN RF EAOUT
Protection
The MAX16821A/MAX16821B/MAX16821C include output overvoltage protection (OVP). During fault conditions when the load goes to high impedance (output opens), the controller attempts to maintain LED current. The OVP disables the MAX16821A/MAX16821B/ MAX16821C whenever the output voltage exceeds the OVP threshold, protecting the external circuits from undesirable voltages.
Figure 4. Overvoltage Protection Input Delay
12
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
Applications Information
Boost LED Driver
Figure 5 shows the MAX16821A/MAX16821B/MAX16821C configured as a synchronous boost converter with MODE connected to VCC. During the on-time, the input voltage charges the inductor. During the off-time, the inductor discharges to the output. The output voltage cannot go below the input voltage in this configuration. Resistor R1 senses the inductor current and resistor R2 senses the LED current. The outer LED current regulation loop programs the average current in the inductor, thus achieving tight LED current regulation.
MAX16821A/MAX16821B/MAX16821C
VCC
VLED
R4 ON/OFF R9 R3 C3 VIN 7V TO 28V
R10 14 C11 15 OVI C10 R8 16 CLP R7 17 EAOUT C9 C8 R5 19 DIFF 18 EAN LX 5 C4 DH 6 Q1 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7 L1
C2
VLED Q2
MAX16821A MAX16821B MAX16821C
BST 4 R5 DL 3
C1 LED STRING R2
20 CSN
N.C. 2
D1
R1
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 5. Synchronous Boost LED Driver (Output Voltage Not to Exceed 28V)
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13
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Input-Referenced Buck-Boost LED Driver
The circuit in Figure 6 shows a step-up/step-down regulator. It is similar to the boost converter in Figure 5 in that the inductor is connected to the input and the MOSFET is essentially connected to ground. However, rather than going from the output to ground, the LEDs span from the output to the input. This effectively removes the boost-only restriction of the regulator in Figure 5, allowing the voltage across the LED to be greater or less than the input voltage. LED currentsensing is not ground-referenced, so a high-side current-sense amplifier is used to measure current.
VCC
VLED
R4 ON/OFF R8 R3 C3 VIN 7V TO 28V R2 C2 LED STRING 1 TO 6 LEDS
C2 R9 14 C11 15 OVI C10 R7 16 CLP R6 17 EAOUT C9 C8 R5 19 DIFF 18 EAN LX 5 C1 DH 6 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7 L1 D1
VLED Q1 VCC
RS+ RSOUT
MAX16821A MAX16821B MAX16821C
BST 4
DL 3
20 CSN
N.C. 2
R1
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 6. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (7V to 28V Input)
14
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
SEPIC LED Driver
Figure 7 shows the MAX16821A/MAX16821B/ MAX16821C configured as a SEPIC LED driver. While buck topologies produce an output always lower than the input, and boost topologies produce an output always greater than the input, a SEPIC topology allows the output voltage to be greater than, equal to, or less than the input. In a SEPIC topology, the voltage across C3 is the same as the input voltage, and L1 and L2 have the same inductance. Therefore, when Q1 turns on (ontime), the currents in both inductors (L1 and L2) ramp up at the same rate. The output capacitor supports the output voltage during this time. When Q1 turns off (offtime), L1 current recharges C3 and combines with L2 to provide current to recharge C1 and supplies the load current. Since the voltage waveform across L1 and L2 are exactly the same, it is possible to wind both inductors on the same core (a coupled inductor). Although voltages on L1 and L2 are the same, RMS currents can be quite different so the windings may require a different gauge wire. Because of the dual inductors and segmented energy transfer, the efficiency of a SEPIC converter is lower than the standard buck or boost configurations. As in the boost driver, the current-sense resistor connects to ground, allowing the output voltage of the LED driver to exceed the rated maximum voltage of the MAX16821A/MAX16821B/MAX16821C.
MAX16821A/MAX16821B/MAX16821C
VCC
VLED
R4 ON/OFF R8 R3 C2 VIN 7V TO 28V
R9 14 C10 15 OVI C9 R7 16 CLP R6 17 EAOUT C8 C7 R5 19 DIFF 18 EAN LX 5 DH 6 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7
L1 C3 D1 VLED
Q1
L2
C1
LED STRING
MAX16821A MAX16821B MAX16821C
BST 4
DL 3
R2
20 CSN
N.C. 2
R1
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C6
C5
C4
Figure 7. Typical Application Circuit for a SEPIC LED Driver
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Low-Side Buck Driver with Synchronous Rectification
In Figure 8, the input voltage goes from 7V to 28V and, because of the ground-based current-sense resistor, the output voltage can be as high as the input. The synchronous MOSFET keeps the power dissipation to a minimum, especially when the input voltage is large compared to the voltage on the LED string. For the inner average current-loop inductor, current is sensed by resistor R1. To regulate the LED current, R2 creates a voltage that the differential amplifier compares to 0.6V. Capacitor C1 is small and helps reduce the ripple current in the LEDs. Omit C1 in cases where the LEDs can tolerate a higher ripple current. The average currentmode control scheme converts the input voltage to a current source feeding the LED string.
VCC R4 ON/OFF R9 R3 C3
VLED
R10 14 C11 15 OVI C10 R9 16 CLP R7 17 EAOUT C9 C8 R6 19 DIFF 18 EAN LX 5 C4 DH 6 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7
VIN 7V TO 28V
C2
Q1 L1 VLED
MAX16821A MAX16821B MAX16821C
R5 BST 4
LED STRING Q2 C1
DL 3
20 CSN
N.C. 2
D2 R1
R2
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 8. Application Circuit for a Low-Side Buck LED Driver
16
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
High-Side Buck Driver with Synchronous Rectification
In Figure 9, the input voltage goes from 7V to 28V, the LED load is connected from the positive side to the currentsense resistor (R1) in series with the inductor, and MODE is connected to VCC. For the inner average current-loop inductor, current is sensed by resistor R1 and is then transferred to the low side by the high-side current-sense amplifier, U2. The voltage appearing across resistor R11 becomes the average inductor current-sense voltage for the inner average current loop. To regulate the LED current, R2 creates a voltage that the differential amplifier compares to its internal reference. Capacitor C1 is small and is added to reduce the ripple current in the LEDs. In cases where the LEDs can tolerate a higher ripple current, capacitor C1 can be omitted.
MAX16821A/MAX16821B/MAX16821C
VCC
R4 ON/OFF C3 R3
14 C11 15 OVI C10 R8 16 CLP R7 17 EAOUT C9 C8 R6 19 DIFF 18 EAN I.C. I.C.
13 OUTV
12 RT/SYNC
11 EN
10 MODE
9 CLKOUT
8 SGND N.C. 7
VIN 7V TO 28V
C2 DH 6 Q1 L1 LX 5 C4 R1 C1
LED STRING
VCC RS+ U2 R2 OUT R11
MAX16821A MAX16821B MAX16821C
R5 BST 4 D1 DL 3
Q2
RS-
20 CSN
N.C. 2
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 9. Application Circuit for a High-Side Buck LED Driver
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17
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Inductor Selection
The switching frequency, peak inductor current, and allowable ripple at the output determine the value and size of the inductor. Selecting higher switching frequencies reduces inductance requirements, but at the cost of efficiency. The charge/discharge cycle of the gate and drain capacitance in the switching MOSFETs create switching losses worsening at higher input voltages, since switching losses are proportional to the square of the input voltage. The MAX16821A/ MAX16821B/MAX16821C operate up to 1.5MHz. Choose inductors from the standard high-current, surface-mount inductor series available from various manufacturers. Particular applications may require custom-made inductors. Use high-frequency core material for custom inductors. High IL causes large peak-topeak flux excursion increasing the core losses at higher frequencies. The high-frequency operation coupled with high IL reduces the required minimum inductance and makes the use of planar inductors possible. The following discussion is for buck or continuous boost-mode topologies. Discontinuous boost, buckboost, and SEPIC topologies are quite different in regards to component selection. Use the following equations to determine the minimum inductance value: Buck regulators: LMIN = Boost regulators: LMIN =
Switching MOSFETs
When choosing a MOSFET for voltage regulators, consider the total gate charge, RDS(ON), power dissipation, and package thermal impedance. The product of the MOSFET gate charge and on-resistance is a figure of merit, with a lower number signifying better performance. Choose MOSFETs optimized for high-frequency switching applications. The average current from the MAX16821A/MAX16821B/MAX16821C gate-drive output is proportional to the total capacitance it drives from DH and DL. The power dissipated in the MAX16821A/MAX16821B/MAX16821C is proportional to the input voltage and the average drive current. The gate charge and drain capacitance losses (CV2), the cross-conduction loss in the upper MOSFET due to finite rise/fall time, and the I2R loss due to RMS current in the MOSFET RDS(ON) account for the total losses in the MOSFET. Estimate the power loss (PDMOS_) in the high-side and low-side MOSFETs using the following equations:
PDMOS _ HI = (QG x VDD x fSW ) + VIN x ILED x (tR + t f ) x fSW + 2
RDSON x I2RMS-HI
(VINMAX
VINMAX x fSW x IL
- VLED ) x VLED
where QG, RDS(ON), tR, and tF are the upper-switching MOSFET's total gate charge, on-resistance, rise time, and fall time, respectively.
IRMS-HI = D I2 VALLEY + I2 PK + I VALLEY x IPK x 3
(VLED
VLED x fSW x IL
- VINMAX ) x VINMAX
For the buck regulator, D is the duty cycle, IVALLEY = (IOUT - IL / 2) and IPK = (IOUT + IL / 2).
PDMOS _ LO = (QG x VDD x fSW ) + RDSON x I2 RMS-LO IRMS-LO =
where VLED is the total voltage across the LED string. The average current-mode control feature of the MAX16821A/MAX16821B/MAX16821C limits the maximum peak inductor current and prevents the inductor from saturating. Choose an inductor with a saturating current greater than the worst-case peak inductor current. Use the following equation to determine the worstcase current in the average current-mode control loop. ILPEAK = VCL I + CL 2 RS
(1- D) I2 VALLEY + I2 PK + I VALLEY x IPK x 3
Input Capacitors
where RS is the sense resistor and VCL = 0.030V. For the buck converter, the sense current is the inductor current and for the boost converter, the sense current is the input current.
18
The discontinuous input-current waveform of the buck converter causes large ripple currents in the input capacitor. The switching frequency, peak inductor current, and the allowable peak-to-peak voltage ripple reflected back to the source dictate the capacitance requirement. The input ripple is comprised of V Q (caused by the capacitor discharge) and V ESR (caused by the ESR of the capacitor).
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High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
Use low-ESR ceramic capacitors with high ripple-current capability at the input. In the case of the boost topology where the inductor is in series with the input, the ripple current in the capacitor is the same as the inductor ripple and the input capacitance is small. Select a 5% lower value of RS to compensate for any parasitics associated with the PCB. Select a non-inductive resistor with the appropriate wattage rating. In the case of the boost configuration, the MAX16821A/ MAX16821B/MAX16821C accurately limits the maximum input current. Use the following equation to calculate the current-sense resistor value: 0.0264 RSENSE = IIN where IIN is the input current.
MAX16821A/MAX16821B/MAX16821C
Output Capacitors
The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and the bulk capacitance of the output capacitor contribute to the output ripple. In most of the applications, the output ESR and ESL effects can be dramatically reduced by using low-ESR ceramic capacitors. To reduce the ESL effects, connect multiple ceramic capacitors in parallel to achieve the required bulk capacitance. In a buck configuration, the output capacitance, COUT, is calculated using the following equation: (VINMAX - VLED ) x VLED COUT VR x 2 x L x VINMAX x fSW 2 where VR is the maximum allowable output ripple. In a boost configuration, the output capacitance, COUT, is calculated as: (VLED - VINMIN ) x 2 x ILED COUT VR x VLED x fSW where ILED is the output current. In a buck-boost configuration, the output capacitance, COUT is: 2 x VLED x ILED COUT VR x (VLED + VINMIN ) x fSW where VLED is the voltage across the load and ILED is the output current.
Compensation
The main control loop consists of an inner current loop (inductor current) and an outer LED current regulation loop. The MAX16821A/MAX16821B/MAX16821C use an average current-mode control scheme to regulate the LED current (Figure 2). The VEA output provides the controlling voltage for the current source. The inner current loop absorbs the inductor pole reducing the order of the LED current loop to that of a single-pole system. The major consideration when designing the current control loop is making certain that the inductor downslope (which becomes an upslope at the output of the CEA) does not exceed the internal ramp slope. This is a necessary condition to avoid subharmonic oscillations similar to those in peak current mode with insufficient slope compensation. This requires that the gain at the output of the CEA be limited based on the following equation: Buck: RCF VRAMP x fSW x L AV x RS x VLED x gm
Average Current Limit
The average current-mode control technique of the MAX16821A/MAX16821B/MAX16821C accurately limits the maximum output current in the case of the buck configuration. The MAX16821A/MAX16821B/MAX16821C sense the voltage across the sense resistor and limit the peak inductor current (IL-PK) accordingly. The on-cycle terminates when the current-sense voltage reaches 26.4mV (min). Use the following equation to calculate the maximum current-sense resistor value: 0.0264 RSENSE = ILED
where VRAMP = 2V, gm = 550S, AV = 34.5V/V, and VLED is the voltage across the LED string. The crossover frequency of the inner current loop is given by: fC = RS VIN x x 34.5 x gm x RCF VRAMP 2x xL
For adequate phase margin place the zero formed by RCF and CCZ at least 3 to 5 times below the crossover frequency. The pole formed by RCF and CCP may not be required in most applications but can be added to minimize noise at a frequency at or above the switching frequency.
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19
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Boost: RCF VRAMP x fSW x L AV x RS x (VLED - VIN ) x gm For adequate phase margin at crossover, place the zero formed by RCF and CCZ at least 3 to 5 times below the crossover frequency. The pole formed by RCF and CCP is added to eliminate noise spikes riding on the current waveform and is placed at the switching frequency.
The crossover frequency of the inner current loop is given by: fC = RS VRAMP x VLED x 34.5 x gm x RCF 2x xL
PWM Dimming
Even though the MAX16821A/MAX16821B/MAX16821C do not have a separate PWM input, PWM dimming can be easily achieved by means of simple external circuitry. See Figures 10 and 11.
VCC R4 ON/OFF R9 R3 C3
VLED
R10 14 C11 15 OVI C10 R9 16 CLP R7 17 EAOUT C9 C8 R6 19 DIFF 18 EAN LX 5 C4 DH 6 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7
VIN 7V TO 28V
C2
Q1 L1 VLED
MAX16821A MAX16821B MAX16821C
R5 BST 4
PWM DIM DL 3 Q2 Q3
LED STRING
20 CSN
N.C. 2
D2 R1
R2
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 10. Low-Side Buck LED Driver with PWM Dimming
20 ______________________________________________________________________________________
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
VCC VLED VCC R8 R10 Q5 PWM DIM Q4 C11 15 OVI C10 R7 16 CLP R6 17 EAOUT C9 PWM DIM Q3 R5 19 DIFF C8 18 EAN LX 5 C1 DH 6 R9 14 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7 Q1 LED STRING L1 D1 VLED R3 C2 C3 R4 ON/OFF VIN 7V TO 28V
MAX16821A MAX16821B MAX16821C
BST 4
PWM DIM Q2
DL 3
R2
20 CSN
N.C. 2
R1
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Figure 11. Boost LED Driver with PWM Dimming
Power Dissipation
Calculate power dissipation in the MAX16821A/ MAX16821B/MAX16821C as a product of the input voltage and the total VCC regulator output current (ICC). ICC includes quiescent current (IQ) and gate-drive current (IDD): PD = VIN x ICC ICC = IQ + [fSW x (QG1 + QG2)]
where QG1 and QG2 are the total gate charge of the low-side and high-side external MOSFETs at VGATE = 5V, IQ is the supply current, and fSW is the switching frequency of the LED driver. Use the following equation to calculate the maximum power dissipation (PDMAX) in the chip at a given ambient temperature (TA): PDMAX = 34.5 x (150 - TA) mW
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21
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
PCB Layout
Use the following guidelines to layout the LED driver. 1) Place the IN, V CC , and V DD bypass capacitors close to the MAX16821A/MAX16821B/MAX16821C. 2) Minimize the area and length of the high-current switching loops. 3) Place the necessary Schottky diodes that are connected across the switching MOSFETs very close to the respective MOSFET. 4) Use separate ground planes on different layers of the PCB for SGND and PGND. Connect both of these planes together at a single point and make this connection under the exposed pad of the MAX16821A/MAX16821B/MAX16821C. 5) Run the current-sense lines CSP and CSN very close to each other to minimize the loop area. Run the sense lines SENSE+ and SENSE- close to each other. Do not cross these critical signal lines with power circuitry. Sense the current right at the pads of the current-sense resistors. The current-sense signal has a maximum amplitude of 27.5mV. To prevent contamination of this signal from high dv/dt and high di/dt components and traces, use a ground plane layer to separate the power traces from this signal trace. 6) Place the bank of output capacitors close to the load. 7) Distribute the power components evenly across the board for proper heat dissipation. 8) Provide enough copper area at and around the switching MOSFETs, inductor, and sense resistors to aid in thermal dissipation. 9) Use 2oz or thicker copper to keep trace inductances and resistances to a minimum. Thicker copper conducts heat more effectively, thereby reducing thermal impedance. Thin copper PCBs compromise efficiency in applications involving high currents.
*EP = EXPOSED PAD.
Selector Guide
PART DIFFERENTIAL SET VALUE (VSENSE+ - VSENSE-) (V) 0.60 0.10 0.03 DIFFERENTIAL AMP GAIN (V/V) 1 6 20
MAX16821A MAX16821B MAX16821C
Pin Configuration
CSN CSP DIFF EAN CLP 16 OVI 15 14 *EP 13 12 I.C. OUTV RT/SYNC EN MODE CLKOUT SGND 11 10 9 8 1 PGND 2 N.C. 3 DL 4 BST 5 LX 6 DH 7 N.C.
TOP VIEW
21 SGND 22 SENSE- 23 SENSE+ 24 SGND 25 IN 26 VCC 27 VDD 28
20
19
18
17
MAX16821A MAX16821B MAX16821C +
TQFN
Chip Information
PROCESS: BiCMOS
22
______________________________________________________________________________________
EAOUT
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing
Typical Operating Circuit
VCC R4 ON/OFF R9 R3 C3
MAX16821A/MAX16821B/MAX16821C
VLED
R10 14 C11 15 OVI C10 R9 16 CLP R7 17 EAOUT C9 C8 R6 19 DIFF 18 EAN LX 5 C4 DH 6 I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND N.C. 7
VIN 7V TO 28V
C2
Q1 L1 VLED
MAX16821A MAX16821B MAX16821C
R5 BST 4
LED STRING Q2 C1
DL 3
20 CSN
N.C. 2
D2 R1
R2
21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 VIN IN 26 VCC 27
PGND 1 VDD 28
C7
C6
C5
Package Information
For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE 28 TQFN-EP PACKAGE CODE T2855-8 DOCUMENT NO. 21-0140
______________________________________________________________________________________
23
High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C
Revision History
REVISION NUMBER 0 1 REVISION DATE 7/07 3/09 Initial release Updated Electrical Characteristics table. DESCRIPTION PAGES CHANGED -- 3, 4
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
24 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.
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